1. Field of the Invention
The present invention relates to an amplifier suitable for a wide band high output circuit such as a driving circuit for, for example, a cathode ray tube display.
2. Description of the Prior Art
Conventionally, a wide band high output video amplifier for driving a cathode ray tube (hereinafter abbreviated as CRT) comprises a bipolar transistor which operates as a driving element and a peaking circuit which compensates for deterioration in the high frequency characteristic in the amplifier due to an input capacitance of the CRT by using an inductance element. FIG. 1 illustrates an example of such a conventional amplifier which will be first described by referring to the drawings. Generally, for a driving circuit in a wide band amplifier, a cascode circuit constituted by a common-emitter stage and a common-base stage (common-gate stage if a field effect transistor (hereinafter abbreviated as FET) is used) is employed.
In FIG. 1, the reference numeral 1 denotes a CRT, 2 denotes a load resistor, 3 denotes a peaking coil, 4 denotes a load capacitor, 5 denotes an output bipolar transistor, 6 denotes a driving bipolar transistor, 7 denotes an emitter resistor, 8 denotes an input signal source, and 9 denotes a bias voltage. In FIG. 1, those elements other than the peaking coil 3 constitute a basic circuit of a cascode video amplifier. Assuming that each of the output and driving bipolar transistors 5 and 6 is sufficiently good in high frequency characteristic, the cut-off frequency of this video amplifier is determined by the factor, that is the time constant (one stage low-pass filter) constituted by the resistance R.sub.L of the load resistor 2 and the capacitance C.sub.L of the load capacitor 4. The capacitance C.sub.L of the load capacitor is the sum of the output capacitance of the transistors 5 and 6, the input capacitance of the CRT 1, and the stray capacitance of wiring and the like, and becomes a main load for the frequency characteristic of the video amplifier in the high frequency range.
The output circuit including the peaking coil 3 as shown in FIG. 1 is called a shunt peaking circuit, in which the load capacitance C.sub.L and the inductance L.sub.P of the peaking coil 3 constitute a parallel resonance circuit to increase the amplification at the upper end of the frequency band so as to expand the band width in the vicinity of the cut-off frequency of the video amplifier by properly adjusting the resonance frequency f.sub.o and the quality factor Q. The frequency characteristic of the load impedance Z.sub.L when the output side is viewed from the output transistor 5 of the shunt peaking circuit illustrated in FIG. 1 may be as shown in FIG. 2. The particular relations among the resonance frequency f.sub.o, the quality factor Q, and the load impedance Z.sub.L, etc. are not described here because they are well known by the teachings of various references.
The bipolar transistors used as amplifying elements in FIG. 1 will be now described. In the case of a cascode amplifier, since the transistor 6 used as an element in the driving stage not only converts a voltage input into a current so as to produce a current output but also operates as a common-emitter (or commonsource) element, it is particularly important that the transistor 6 has a good high-frequency-characteristic and a large mutual conductance gm. Since the maximum voltage applied to the driving transistor 5 is made to be the bias voltage, the driving transistor 6 may be selected to have relatively low breakdown voltage. That is, generally, a high frequency transistor of low breakdown voltage can be used for the element in the driving stage. This is helpful in view of the technique for producing high frequency bipolar transistors so that the element in the driving stage is easily available and there is a large degree of freedom for selecting the element. The transistor in the output stage is, on the other hand, used with the common-base (or common-gate) and, therefore, it may be used to the utmost limit of the capability in its high frequency characteristic. Since the transistor in the output stage is, however, required to satisfy various contradictory requests such as a large allowable collector power dissipation with high breakdown voltage, a small output capacitance, a good high frequency characteristic, or the like, it has a difficult or serious problem in selection and design thereof. Particularly, in recent experiments, an important problem has appeared that the collector volume resistance increases to cause a false saturation phenomenon due to a so-called emitter fringing effect, that is the emitter region becomes substantially narrow under the condition of a high frequency large current in spite of the fact that the element has a cut-off frequency to make it possible to sufficiently cover the used band. The false saturation phenomenon causes the collector volume resistance to increase to thereby reduce the effective output current to restrict the possible maximum power output. This leads not only to the necessity of increasing the source voltage to thereby cause an increase in power dissipation in the collector and in power consumption in the circuit but also the need for an element having higher breakdown voltage, resulting in a problem that the compatibility with the high frequency characteristic becomes more difficult.
Thus, it has been found that there is a serious problem to be solved in the case where a bipolar transistor is used in the output stage. On the other hand, it has become apparent that the power MOS FET which has been recently initiated for use in audio instruments or switching regulators has very superior performance also in use for video output. Particularly, for the output element in the cascode amplifier in FIG. 1, the power MOS FET has very superior merits as follows:
(1) It is easy to make the breakdown voltage high without deteriorating the high frequency characteristic to such an extent;
(2) The area of safe operation (ASO) is wide; and
(3) No false saturation phenomenon occurs, etc. The power MOS FET has, on the other hand, a very serious structural defect in the high frequency range in that the drain zone and the source zone are connected to each other through capacitance. The difference in structure between the FET and the bipolar transistor will be briefly described by referring to FIGS. 3 and 4.
Although the description will be made with the most popular power MOS FET as a model of FET in this application, the discussion of course applies to the static induction transistor, or the like.
FIGS. 3A and 3B illustrate the general structure of a bipolar transistor and an equivalent circuit thereof when the bipolar transistor is used with the common-base. In FIGS. 3A and 3B, various reference characters are used to denote various parameters as follows:
E: an emitter, B: a base, C: a collector, C.sub.be : base-emitter capacitance, r.sub.be : base-emitter resistance, v.sub.be : base-emitter voltage, gm: mutual conductance, C.sub.ob : output capacitance of collector. Although further parameters are required to describe in detail, the characteristics of a bipolar transistor in video frequency band may be approximated by those parameters used in FIGS. 3A and 3B. The most important characteristic of the bipolar transistor as shown in FIG. 3 is that the mutual conductance gm is large and the collector-emitter capacitance is negligible small.
Next, the power MOS FET as shown in FIGS. 4A and 4B will be described. In FIGS. 4A and 4B, the same reference characters as those already used in FIGS. 3A and 3B denote the same parameters and other reference characters are used to denote the parameters as follows:
S: a source, G: a gate, D: a drain, C.sub.gs : gate-source capacitance, C.sub.sd : source-drain capacitance, C.sub.dg : drain-gate capacitance, v.sub.gs : gate-source voltage.
As will be apparent from FIGS. 3A, 3B, 4A and 4B, in the case of the bipolar transistor only the current source gm.multidot.v.sub.be is inserted between the collector and emitter, while in the case of the power MOS FET the source-drain capacitance C.sub.sd is additionally inserted in parallel with the current source gm.multidot.v.sub.gs between the drain and source so that it may be considered that the source and the drain are completely connected through the sourcedrain capacitance C.sub.sd in the sense of high frequency, resulting in a very important defect of the MOS FET as will be described later.
Next, discussion will be made as to the input impedance characteristic of each of a bipolar transistor and a MOS FET, in the cases of the common-base and common-gate. The relation between the absolute value of the input impedance and the frequency of the bipolar transistor and the power MOS FET is as shown by solid line in FIGS. 5 and 6 respectively. In the case of the bipolar transistor of FIG. 5, the absolute value of the input impedance monotonously changes as the frequency changes, while in the case of the power MOS FET in FIG. 6, the absolute value of the input impedance has a peak at a certain frequency (shown by f' in FIG. 6) mainly due to the relation between the mutual conductance gm and the source-drain capacitance C.sub.sd. If calculation is made by using the simple equivalent circuit as shown in FIG. 4 (a relatively good approximation can be obtained in the case of a MOS FET having a small gate resistance), the peak of the absolute value .vertline.Z.sub.in .vertline. of the input impedance Z.sub.in becomes smaller and the peak frequency moves toward the higher frequency zone as the mutual conductance gm becomes larger under the condition that the source-drain capacitance C.sub.sd is fixed. Alternatively, if the source-drain capacitance C.sub.sd is caused to increase with the mutual conductance gm fixed, the peak of the absolute value .vertline.Z.sub.in .vertline. increases and the peak frequency moves toward the lower frequency zone. In the case where a power MOS FET is used in a video output stage, it is necessary to make the peak value .vertline.Z.sub.in .vertline. sufficiently small or make the peak frequency sufficiently small or make the peak frequency sufficiently high and to select one having a substantially flat characteristic over the used frequency band.
Next, the characteristic shown by dotted line in FIGS. 5 and 6 will be described. The input impedance characteristic in the cases of the common-base and the common-gate are shown by dotted line in FIGS. 5 and 6 respectively when an impedance element having an impedance value which has a peak at a certain frequency. In the case of a power MOS FET as shown in FIG. 6, the effect of the drain side is apparent in that the peak value of .vertline.Z.sub.in .vertline. becomes maximum when f.sub.o coincides with f'. This arises a very serious problem when a power MOS FET is used as an output transistor in such a cascode amplifier as shown in FIG. 1. That is, in the common-gate element which has its remarkable characteristic in its low input impedance, if the input impedance of an output transistor takes a very large value at a certain frequency, a large voltage amplitude is produced at the collector of the driving bipolar transistor in FIG. 1 and the frequency characteristic is deteriorated due to Miller effect at the base side of the driving bipolar transistor. Further, the bias voltage E.sub. B is generally selected to be about 10 volts in the viewpoint of the collector power dissipation and the breakdown voltage in the driving transistor and there is much possibility to cause the worst situation in which the output waveform is clipped due to shortage of the dynamic range of the driving bipolar transistor in case where a large voltage amplitude is produced at the collector of the driving bipolar transistor.
As described above, a power MOS FET having a remarkable merit in comparison with a bipolar transistor when used as an output element in a wide high output amplifier, has also a serious defect so that it is necessary to pay sufficient attention in using the power MOS FET and the scope of application of the power MOS FET is very narrow.